Pulsed level shift and inverter circuits for gan devices

ABSTRACT

GaN-based half bridge power conversion circuits employ control, support and logic functions that are monolithically integrated on the same devices as the power transistors. In some embodiments a low side GaN device communicates through one or more level shift circuits with a high side GaN device. Various embodiments of level shift circuits and their inventive aspects are disclosed.

CROSS REFERENCE TO RELATED APPLICATIONS

This application claims priority to and is a continuation of U.S. patentapplication Ser. No. 16/699,081, for PULSED LEVEL SHIFT AND INVERTERCIRCUITS FOR GAN DEVICES, filed on Nov. 28, 2019, which is a divisionalof U.S. patent application Ser. No. 16/151,695, for “PULSED LEVEL SHIFTAND INVERTER CIRCUITS FOR GAN DEVICES” filed on Oct. 4, 2018, now U.S.Pat. No. 10,530,169, issued on Jan. 7, 2020, which is a divisional ofU.S. patent application Ser. No. 15/219,248, for “PULSED LEVEL SHIFT ANDINVERTER CIRCUITS FOR GAN DEVICES” filed on Jul. 25, 2016, now U.S. Pat.No. 10,135,275, issued on Nov. 20, 2018, which is a continuation of U.S.patent application Ser. No. 14/667,523, for “PULSED LEVEL SHIFT ANDINVERTER CIRCUITS FOR GAN DEVICES” filed on Mar. 24, 2015, now U.S. Pat.No. 9,401,612, issued on Jul. 26, 2016, which claims priority to U.S.provisional patent application No. 62/127,725, for “HALF BRIDGE POWERCONVERSION CIRCUITS USING GAN AND SILICON DEVICES” filed on Mar. 3,2015, and also claims priority to U.S. provisional patent applicationNo. 62/051,160, for “HYBRID HALF-BRIDGE DRIVER USING GAN AND SILICONDEVICES” filed on Sep. 16, 2014. All of the aforementioned disclosuresare hereby incorporated by reference in their entirety for all purposes.

FIELD

The present invention relates generally to power conversion circuits andin particular to power conversion circuits utilizing one or moreGaN-based semiconductor devices.

BACKGROUND BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a simplified schematic of a half bridge power conversioncircuit according to an embodiment of the invention;

FIG. 2 is a simplified schematic of the circuits within the low sidecontrol circuit illustrated in FIG. 1 ;

FIG. 3 is a schematic of the first level shift transistor illustrated inFIG. 1 ;

FIG. 4 is a schematic of the level shift driver circuit illustrated inFIG. 1 ;

FIG. 5 is a schematic of the blanking pulse generator circuitillustrated in FIG. 1 ;

FIG. 6 is an example of waveforms within the blanking pulse generatorillustrated in FIG. 5 ;

FIG. 7 is a schematic of the bootstrap transistor drive circuitillustrated in FIG. 1 ;

FIG. 8 is a block diagram for the low side transistor drive circuitillustrated in FIG. 1

FIG. 9 is a schematic of the startup circuit illustrated in FIG. 1 ;

FIG. 10 is series of diode connected GaN-based enhancement-modetransistors that may be used as a diode clamp in the schematic of FIG. 9;

FIG. 11 is a schematic of the UVLO circuit illustrated in FIG. 1 ;

FIG. 12 is a schematic of the bootstrap capacitor charging circuitillustrated in FIG. 1 ;

FIG. 13 is a schematic of an alternative bootstrap capacitor chargingcircuit as compared to the circuit illustrated in FIG. 12 ;

FIG. 14 is a schematic of the high side logic and control circuitillustrated in FIG. 1 ;

FIG. 15 is a schematic of the first level shift receiver circuitillustrated in FIG. 14 ;

FIG. 16 is a schematic of the second level shift receiver circuitillustrated in FIG. 14 ;

FIG. 17 is a schematic of the pull up trigger circuit illustrated inFIG. 14 ;

FIG. 18 is a schematic of the high side UVLO circuit illustrated in FIG.14 ;

FIG. 19 is a schematic of the high side transistor driver circuitillustrated in FIG. 14 ;

FIG. 20 is a schematic of a high side reference voltage generationcircuit illustrated in FIG. 14 ;

FIG. 21 is a simplified schematic of a half bridge power conversioncircuit according to another embodiment of the invention;

FIG. 22 is a simplified schematic of the circuits within the low sidecontrol circuit illustrated in FIG. 21 ;

FIG. 23 is a schematic of the first level shift transistor illustratedin FIG. 22 ;

FIG. 24 is a schematic of the inverter/buffer circuit illustrated inFIG. 22 ;

FIG. 25 is a schematic of the on pulse generator circuit illustrated inFIG. 22 ;

FIG. 26 is a schematic of the off pulse generator circuit illustrated inFIG. 22 ;

FIG. 27 is a schematic of the blanking pulse generator circuitillustrated in FIG. 22 ;

FIG. 28 is a schematic of the low side transistor drive circuitillustrated in FIG. 22 ;

FIG. 29 is a simplified schematic of the circuits within the high sidecontrol circuit illustrated in FIG. 21 ;

FIG. 30 is a schematic of the level shift 1 receiver circuit illustratedin FIG. 29 ;

FIG. 31 is a schematic of level shift 2 receiver circuit illustrated inFIG. 29 ;

FIG. 32 is a schematic of the high side UVLO circuit illustrated in FIG.29 ;

FIG. 33 is a schematic of the high side transistor driver circuitillustrated in FIG. 29 ;

FIG. 34 is a schematic of an electro-static discharge (ESD) clampcircuit according to an embodiment of the invention;

FIG. 35 is a schematic of an electro-static discharge (ESD) clampcircuit according to an embodiment of the invention;

FIG. 36 is an illustration of a portion of an electronic packageaccording to an embodiment of the invention; and

FIG. 37 is an illustration of the electronic package of FIG. 36 .

DETAILED DESCRIPTION

Certain embodiments of the present invention relate to half bridge powerconversion circuits that employ one or more gallium nitride (GaN)devices. While the present invention can be useful for a wide variety ofhalf bridge circuits, some embodiments of the invention are particularlyuseful for half bridge circuits designed to operate at high frequenciesand/or high efficiencies with integrated driver circuits, integratedlevel shift circuits, integrated bootstrap capacitor charging circuits,integrated startup circuits and/or hybrid solutions using GaN andsilicon devices, as described in more detail below.

Half Bridge Circuit #1

Now referring to FIG. 1 , in some embodiments circuit 100 may include apair of complementary power transistors (also referred to herein asswitches) that are controlled by one or more control circuits configuredto regulate power delivered to a load. In some embodiments a high sidepower transistor is disposed on a high side device along with a portionof the control circuit and a low side power transistor is disposed on alow side device along with a portion of the control circuit, asdescribed in more detail below.

The integrated half bridge power conversion circuit 100 illustrated inFIG. 1 includes a low side GaN device 103, a high side GaN device 105 aload 107, a bootstrap capacitor 110 and other circuit elements, asillustrated and discussed in more detail below. Some embodiments mayalso have an external controller (not shown in FIG. 1 ) providing one ormore inputs to circuit 100 to regulate the operation of the circuit.Circuit 100 is for illustrative purposes only and other variants andconfigurations are within the scope of this disclosure.

In one embodiment, low side GaN device 103 may have a GaN-based low sidecircuit 104 that includes a low side power transistor 115 having a lowside control gate 117. Low side circuit 104 may further include anintegrated low side transistor driver 120 having an output 123 connectedto low side transistor control gate 117. In another embodiment high,side GaN device 105 may have a GaN-based high side circuit 106 thatincludes a high side power transistor 125 having a high side controlgate 127. High side circuit 106 may further include an integrated highside transistor driver 130 having an output 133 connected to high sidetransistor control gate 127.

A voltage source 135 (also known as a rail voltage) may be connected toa drain 137 of high side transistor 125, and the high side transistormay be used to control power input into power conversion circuit 100.High side transistor 125 may further have a source 140 that is coupledto a drain 143 of low side transistor 115, forming a switch node 145.Low side transistor 115 may have a source 147 connected to ground. Inone embodiment, low side transistor 115 and high side transistor 125 maybe GaN-based enhancement-mode field effect transistors. In otherembodiments low side transistor 115 and high side transistor 125 may beany other type of device including, but not limited to, GaN-baseddepletion-mode transistors, GaN-based depletion-mode transistorsconnected in series with silicon based enhancement-mode field-effecttransistors having the gate of the depletion-mode transistor connectedto the source of the silicon-based enhancement-mode transistor, siliconcarbide based transistors or silicon-based transistors.

In some embodiments high side device 105 and low side device 103 may bemade from a GaN-based material. In one embodiment the GaN-based materialmay include a layer of GaN on a layer of silicon. In further embodimentsthe GaN based material may include, but not limited to, a layer of GaNon a layer of silicon carbide, sapphire or aluminum nitride. In oneembodiment the GaN based layer may include, but not limited to, acomposite stack of other III nitrides such as aluminum nitride andindium nitride and III nitride alloys such as AlGaN and InGaN. Infurther embodiments, GaN-based low side circuit 104 and GaN-based highside circuit 106 may be disposed on a monolithic GaN-based device. Inother embodiments GaN-based low side circuit 104 may be disposed on afirst GaN-based device and GaN-based high side circuit 106 may bedisposed on a second GaN-based device. In yet further embodimentsGaN-based low side circuit 104 and GaN-based high side circuit 106 maybe disposed on more than two GaN-based devices. In one embodiment,GaN-based low side circuit 104 and GaN-based high side circuit 106 maycontain any number of active or passive circuit elements arranged in anyconfiguration.

Low Side Device

Low side device 103 may include numerous circuits used for the controland operation of the low side device and high side device 105. In someembodiments, low side device 103 may include logic, control and levelshift circuits (low side control circuit) 150 that controls theswitching of low side transistor 115 and high side transistor 125 alongwith other functions, as discussed in more detail below. Low side device103 may also include a startup circuit 155, a bootstrap capacitorcharging circuit 157 and a shield capacitor 160, as also discussed inmore detail below.

Now referring to FIG. 2 , the circuits within low side control circuit150 are functionally illustrated. Each circuit within low side controlcircuit 150 is discussed below, and in some cases is shown in moredetail in FIGS. 3-14 . In one embodiment the primary function of lowside control circuit 150 may be to receive one or more input signals,such as a PWM signal from a controller, and control the operation of lowside transistor 115, and high side transistor 125.

In one embodiment, first and a second level shift transistors 203, 205,respectively, may be employed to communicate with high side logic andcontrol circuit 153 (see FIG. 1 ). In some embodiments, first levelshift transistor 203 may be a high voltage enhancement-mode GaNtransistor. In further embodiments, first level shift transistor 203 maybe similar to low side transistor 115 (see FIG. 1 ) and high sidetransistor 125, except it may be much smaller in size (e.g., first levelshift transistor may be tens of microns in gate width with minimumchannel length).

In other embodiments first level shift transistor 203 may experiencehigh voltage and high current at the same time (i.e. the device mayoperate at the high power portion of the device Safe Operating Area) foras long as high side transistor 125 (see FIG. 1 ) is on. Such conditionsmay cause relatively high power dissipation, thus some embodiments mayinvolve design and device reliability considerations in the design offirst level shift transistor 203, as discussed in more detail below. Infurther embodiments, a first level shift resistor 207 may be added inseries with a source 210 of first level shift transistor 203 to limitgate 213 to source 210 voltage and consequently the maximum currentthrough the first level shift transistor. Other methods may be employedto limit the current through first level shift transistor 203, and arewithin the scope of this disclosure. Drain 215 of first level shifttransistor 203 may be coupled to high side logic and control circuit 153(see FIG. 1 ), as discussed in more detail below.

In one embodiment, first level shift transistor 203 may comprise aportion of an inverter circuit having a first input and a first outputand configured to receive a first input logic signal at the first inputterminal and in response, provide a first inverted output logic signalat the first output terminal, as discussed in more detail below. Infurther embodiments the first input and the first inverted output logicsignals can be referenced to different voltage potentials. In someembodiments, first level shift resistor 207 may be capable of operatingwith the first inverted output logic signal referenced to a voltage thatis more than 13 volts higher than a reference voltage for the firstinput logic signal. In other embodiments it may be capable of operatingwith the first inverted output logic signal referenced to a voltage thatis more than 20 volts higher than a reference voltage for the firstinput logic signal, while in other embodiments it may be between 80-400volts higher.

In other embodiments, first level shift resistor 207 may be replaced byany form of a current sink. For example, in one embodiment, source 210of first level shift transistor 203 may be connected to a gate to sourceshorted depletion-mode device. In a further embodiment, thedepletion-mode device may be fabricated by replacing theenhancement-mode gate stack with a high voltage field plate metalsuperimposed on top of the field dielectric layers. The thickness of thefield dielectric and the work function of the metal may be used todetermine the pinch-off voltage of the stack.

In other embodiments first level shift resistor 207 may be replaced by acurrent sink. The current sink may use a reference current (Iref) thatmay be generated by startup circuit 155 (illustrated in FIG. 1 anddiscussed in more detail below). Both the depletion-mode transistor andcurrent sink embodiments may result in a significant device areareduction compared to the resistor embodiment (i.e., because arelatively small depletion-mode transistor would suffice and Iref isalready available from startup circuit 155).

Second level shift transistor 205 may be designed similar to first levelshift transistor 203 (e.g., in terms of voltage capability, currenthandling capability, thermal resistance, etc.). Second level shifttransistor 205 may also be built with either an active current sink or aresistor, similar to first level shift transistor 203. In one embodimentthe primary difference with second level shift transistor 205 may be inits operation. In some embodiments the primary purpose of second levelshift transistor 205 may be to prevent false triggering of high sidetransistor 125 (see FIG. 1 ) when low side transistor 115 turns off.

In one embodiment, for example, false triggering can occur in a boostoperation when low side transistor 115 turn off results in the loadcurrent flowing through high side transistor 125 while the transistor isoperating in the third quadrant with its gate shorted to its source(i.e., in synchronous rectification mode). This condition may introducea dv/dt condition at switch node (Vsw) 145 since the switch node was ata voltage close to ground when low side transistor 115 was on and thentransitions to rail voltage 135 over a relatively short time period. Theresultant parasitic C*dv/dt current (i.e., where C=Coss of first levelshift transistor 203 plus any other capacitance to ground) can causefirst level shift node 305 (see FIG. 3 ) to get pulled low which willthen turn on high side transistor 125. In some embodiments thiscondition may not be desirable because there may be no dead timecontrol, and shoot through may occur from high side transistor 125 andlow side transistor 115 being in a conductive state simultaneously.

FIG. 3 illustrates one embodiment showing how first level shifttransistor 203 may be electrically coupled to high side device 105.First level shift transistor 203, located on low side device 103, isillustrated along with a pull up resistor 303 that may be located onhigh side device 105 (see FIG. 1 ). In some embodiments, first levelshift transistor 203 may operate as a pull down transistor in a resistorpull up inverter.

In further embodiments, when level shift driver circuit 217 (see FIG. 2) supplies a high gate signal (L1_DR) to first level shift transistor203, a first level shift node 305 gets pulled low which is inverted byhigh side logic and control circuit 153 (see FIG. 1 ). The invertedsignal appears as a high state signal that turns on high side transistor137 (see FIG. 1 ) which then pulls the voltage at switch node (Vsw) 145close to rail voltage 135.

Conversely, when level shift driver circuit 217 (see FIG. 2 ) supplies alow gate signal to first level shift transistor 203, a first level shiftnode 305 gets pulled to a high logic state which is inverted by highside logic and control circuit 153 (see FIG. 1 ). The inverted signalappears as a low logic state signal that turns off high side transistor125. This scheme may result in a non-inverted gate signal to high sidetransistor 125. In further embodiments, first level shift transistor 203may be designed large enough to be able to pull down on first levelshift node 305, but not so large that its drain to source and drain tosubstrate (i.e., the semiconductor substrate) capacitances induce falsetriggering of high side logic and control circuit 153.

In some embodiments pull up resistor 303 may instead be anenhancement-mode transistor, a depletion-mode transistor or a referencecurrent source element. In further embodiments pull up resistor 303 maybe coupled between the drain and the positive terminal of a floatingsupply (e.g., a bootstrap capacitor, discussed in more detail below)that is referenced to a different voltage rail than ground. In yetfurther embodiments there may be a first capacitance between the firstoutput terminal (LS NODE) 305 and switch node (Vsw) 145 (see FIG. 1 )and a second capacitance between the first output terminal and ground,where the first capacitance is greater than the second capacitance. Thefirst capacitance may be designed such that in response to a high dv/dtsignal at switch node (Vsw) 145 (see FIG. 1 ), a large portion of theC*dv/dt current is allowed to conduct through the first capacitanceensuring that the voltage at first output terminal 305 tracks thevoltage at the switch node (Vsw). In some embodiments shield capacitor160 (see FIG. 1 ) may be designed to act as the first capacitor asdescribed above. In further embodiments shield capacitor 160 (see FIG. 1) may be used to create capacitance between first output terminal 305and switch node (Vsw) 145 (see FIG. 1 ) in half bridge power conversioncircuit 100. In yet further embodiments, shield capacitor 160 (see FIG.1 ) may also be used to minimize a capacitance between first outputterminal 305 and substrate (i.e., the semiconductor substrate). Morespecifically, in some embodiments shield capacitor 160 may be created byadding a conductive shield layer to the device and coupling the layer toswitch node (Vsw) 145. This structure may effectively create twocapacitors. One capacitor is coupled between output terminal 305 andswitch node (Vsw) 145, and the other is coupled between the switch nodeand the substrate. The capacitance between output terminal 305 and thesubstrate is thereby practically eliminated. In further embodimentsshield capacitor 160 (see FIG. 1 ) may be constructed on the low sidechip 103.

Logic, control and level shifting circuit 150 (see FIG. 2 ) may haveother functions and circuits such as, but not limited to, a level shiftdriver circuit 217, a low side transistor drive circuit 120, a blankingpulse generator 223, a bootstrap transistor drive circuit 225 and anunder voltage lock out (UVLO) circuit 227, as explained in separatefigures with more detail below.

Now referring to FIG. 4 , level shift driver circuit 217 is shown ingreater detail. In one embodiment level shift driver circuit 217 mayinclude a first inverter 405 and a second inverter 410 in a sequentialchain. In further embodiments, since level shift driver circuit 217 maybe driving a small gate width first level shift transistor 203, theremay be no need for a buffer stage.

In one embodiment, level shift driver circuit 217 is driven directly bythe pulse-width modulated high side signal (PWM_HS) from the controller(not shown). In some embodiments the (PWM_HS) signal may be supplied byan external control circuit. In one embodiment the external controlcircuit may be an external controller that is in the same package withhigh side device 105, low side device 103, both devices, or packaged onits own. In further embodiments, level shift driver circuit 217 may alsoinclude logic that controls when the level shift driver circuitcommunicates with first level shift transistor 203 (see FIG. 3 ). In oneembodiment an optional low side under voltage lock out signal (LS_UVLO)may be generated by an under voltage lock out circuit within level shiftdriver circuit 217. The low side under voltage lock out circuit can beused to turn off level shift driver circuit 217 if either (Vcc) or (Vdd)for the low side (Vdd_LS) go below a certain reference voltage, or afraction of the reference voltage.

In further embodiments level shift driver circuit 217 may generate ashoot through protection signal for the low side transistor (STP_LS)that is used to prevent shoot through arising from overlapping gatesignals on low side transistor 115 and high side transistor 125. Thefunction of the (STP_LS) signal may be to ensure that low side drivercircuit 120 (see FIG. 2 ) only communicates with the gate terminal ofthe low side transistor 115 when the gate signal to high side transistor125 is low. In other embodiments, the output of first inverter 405 maybe used to generate the shoot through protection signal (STP_LS) for thelow side transistor 115.

In further embodiments, logic for UVLO and shoot-through protection mayimplemented by adding a multiple input NAND gate to first inverter 405,where the inputs to the NAND gate are the (PWM_HS), (LS_UVLO) and(STP_HS) signals. In yet further embodiments, first inverter 405 mayonly respond to the (PWM_HS) signal if both (STP_HS) and (LS_UVLO)signals are high. In further embodiments, the STP_HS signal may begenerated from the low side gate driver block 120, as explained inseparate figures with more detail.

Now referring to FIG. 5 , blanking pulse generator 223 may be used togenerate a pulse signal that corresponds to the turn off transient oflow side transistor 115. This pulse signal may then turn on second levelshift transistor 205 for the duration of the pulse, which triggers acontrol circuit on high side device 105 (see FIG. 1 ) to prevent falsepull down of first level shift node 305 voltage.

FIG. 5 illustrates a schematic of one embodiment of blanking pulsegenerator 223. In some embodiments a low side transistor 115 gate signal(LS_GATE) is fed as an input to blanking pulse generator 223. The(LS_GATE) signal is inverted by a first stage inverter 505, then sentthrough an RC pulse generator 510 to generate a positive pulse. In someembodiments an inverted signal may be needed because the pulsecorresponds to the falling edge of the (LS_GATE) signal. A capacitor 515in RC pulse generator 510 circuit may be used as a high pass filterallowing the dv/dt at its input to appear across resistor 520. Once thedv/dt vanishes at the input to the RC pulse generator 510, capacitor 515may charge slowly through resistor 520, resulting in a slow decayingvoltage waveform across the resistor. The pulse may then be sent througha second inverter 525, a third inverter 530 and a buffer 535 to generatea square wave pulse for the blanking pulse (B_PULSE) signal. Theduration of the pulse may be determined by the value of capacitor 515and resistor 520 in RC pulse generator 510. In some embodiments,capacitor 515 may be constructed using a drain to source shortedenhancement-mode GaN transistor.

Now referring to FIG. 6 , example waveforms 600 within blanking pulsegenerator 223 are illustrated for one embodiment. Trace 605 shows afalling edge of the low side gate pulse (LS_GATE). Trace 610 shows therising edge of first stage inverter 505 output. Trace 615 shows theoutput of RC pulse generator 510 and trace 620 shows the resultingblanking pulse (B_PULSE) signal that is an output of blanking pulsegenerator 223.

Now referring to FIG. 7 , bootstrap transistor drive circuit 225 isillustrated in greater detail. Bootstrap transistor drive circuit 225includes inverter 730, first buffer 735 and second buffer 745. Bootstraptransistor drive circuit 225 may receive the (BOOTFET_DR_IN) signal fromlow side driver circuit 120. The (BOOTFET_DR_IN) signal may be invertedwith respect to the LS_GATE signal. Bootstrap transistor drive circuit225 may be configured to provide a gate drive signal called (BOOTFET_DR)to a bootstrap transistor in bootstrap charging circuit 157 (see FIG. 1), discussed in more detail below. The (BOOTFET_DR) gate drive signalmay be timed to turn on the bootstrap transistor when low sidetransistor 115 is turned on. Also, since bootstrap transistor drivecircuit 225 is driven by (Vcc), the output of this circuit may have avoltage that goes from 0 volts in a low state to (Vcc)+6 volts in a highstate. In one embodiment the bootstrap transistor is turned on after lowside transistor 115 is turned on, and the bootstrap transistor is turnedoff before the low side transistor is turned off.

In some embodiments, the turn on transient of the (BOOTFET_DR) signalmay be delayed by the introduction of a series delay resistor 705 to theinput of second buffer 745, that may be a gate of a transistor in afinal buffer stage. In further embodiments, the turn off transient oflow side transistor 115 (see FIG. 1 ) may be delayed by the addition ofa series resistor to a gate of a final pull down transistor in low sidedrive circuit 120. In one embodiment, one or more capacitors may be usedin bootstrap transistor drive circuit 225, and support voltages of theorder of (Vcc) which, for example, could be 20 volts, depending on theend user requirements and the design of the circuit. In some embodimentsthe one or more capacitors may be made with a field dielectric to GaNcapacitor instead of a drain to source shorted enhancement-modetransistor.

Now referring to FIG. 8 a block diagram for low side transistor drivecircuit 120 is illustrated. Low side transistor drive circuit 120 mayhave a first inverter 805, a buffer 810, a second inverter 815, a secondbuffer 820 and a third buffer 825. Third buffer 825 may provide the(LS_GATE) signal to low side transistor 115 (see FIG. 1 ). In someembodiments two inverter/buffer stages may be used because the input tothe gate of low side transistor 115 (see FIG. 1 ) may be synchronouswith (Vin). Thus, (Vin) in a high state may correspond to (Vgate) of lowside transistor 115 in a high state and vice versa.

In further embodiments, certain portions of low side drive circuit 120may have an asymmetric hysteresis. Some embodiments may includeasymmetric hysteresis using a resistor divider 840 with a transistorpull down 850.

Further embodiments may have multiple input NAND gates for the (STP_LS)signal (shoot through protection on low side transistor 115). In oneembodiment, low side drive circuit 120 may receive the shoot throughprotection signal (STP_LS) from level shift driver circuit 217. Thepurpose of the (STP_LS) signal may be similar to the (STP_HS) signaldescribed previously. The (STP_LS) signal may ensure that low sidetransistor drive circuit 120 does not communicate with gate 117 (seeFIG. 1 ) of low side transistor 115 when level shift driver circuit 217output is at a high state. In other embodiments, the output of the firstinverter stage 805 may be used as the (STP_HS) signal for level shiftdrive circuit 217 and the (BOOTFET_DR_IN) signal for bootstraptransistor drive circuit 225.

In some embodiments, low side transistor drive circuit 120 may employmultiple input NAND gates for the (LS_UVLO) signal received from UVLOcircuit 227 (see FIG. 2 ). Further embodiments may employ a turn offdelay resistor that may be in series with a gate of a final pull downtransistor in final buffer stage 825. The delay resistor may be used insome embodiments to make sure the bootstrap transistor is turned offbefore low side transistor 115 turns off.

Now referring to FIG. 9 , startup circuit 155 is illustrated in greaterdetail. Startup circuit 155 may be designed to have a multitude offunctionalities as discussed in more detail below. Primarily, startupcircuit 155 may be used to provide an internal voltage (in this caseSTART_Vcc) and provide enough current to support the circuits that arebeing driven by (Vcc). This voltage may remain on to support thecircuits until (Vcc) is charged up to the required voltage externallyfrom rail voltage 135 (V+). Startup circuit 155 may also provide areference voltage (Vref) that may be independent of the startup voltage,and a reference current sink (Iref).

In one embodiment, a depletion-mode transistor 905 may act as theprimary current source in the circuit. In further embodimentsdepletion-mode transistor 905 may be formed by a metal layer disposedover a passivation layer. In some embodiments, depletion-mode transistor905 may use a high voltage field plate (typically intrinsic to anyhigh-voltage GaN technology) as the gate metal. In further embodiments afield dielectric may act as the gate insulator. The resultant gatedtransistor may be a depletion-mode device with a high channel pinch-offvoltage (Vpinch) (i.e., pinch-off voltage is proportional to the fielddielectric thickness). Depletion-mode transistor 905 may be designed toblock relatively high voltages between its drain (connected to V+) andits source. Such a connection may be known as a source followerconnection. Depletion-mode transistor 905 may have a gate 906 coupled toground, a source 907 coupled to a first node 911 and a drain 909 coupledto voltage source 135.

In further embodiments a series of identical diode connectedenhancement-mode low-voltage transistors 910 may be in series withdepletion-mode transistor 905. Series of identical diode connectedenhancement-mode low-voltage transistors 910 may be connected in seriesbetween a first node 911 and a second node 912. One or more intermediatenodes 913 may be disposed between each of series of identical diodeconnected enhancement-mode low-voltage transistors 910. The width tolength ratio of the transistors may set the current drawn from (V+) aswell as the voltage across each diode. To remove threshold voltage andprocess variation sensitivity, series of identical diode connectedenhancement-mode low-voltage transistors 910 may be designed as largechannel length devices. In some embodiments, series of identical diodeconnected enhancement-mode low-voltage transistors 910 may be replacedwith one or more high value resistors.

In further embodiments, at the bottom end of series of identical diodeconnected enhancement-mode low-voltage transistors 910, a current mirror915 may be constructed from two enhancement-mode low-voltage transistorsand used to generate a reference current sink (Iref). First currentmirror transistor 920 may be diode connected and second current mirrortransistor 925 may have a gate connected to the gate of the firstcurrent mirror transistor. The sources of first and second currentmirror transistors 920, 925, respectively may be coupled and tied toground. A drain terminal of first current mirror transistor 920 may becoupled to second junction 912 and a source terminal of second currentmirror transistor 925 may be used as a current sink terminal. This stackof current mirror 915 and series of identical diode connectedenhancement-mode low-voltage transistors 910 may form what is known as a“source follower load” to depletion-mode transistor 905.

In other embodiments, when gate 906 of depletion-mode transistor 905 istied to ground, source 907 of the depletion-mode transistor may assume avoltage close to (Vpinch) when current is supplied to the “sourcefollower load”. At the same time the voltage drop across diode connectedtransistor 920 in current mirror 915 may be close to the thresholdvoltage of the transistor (Vth). This condition implies that the voltagedrop across each of series of identical diode connected enhancement-modelow-voltage transistors 910 may be equal to (Vpinch−Vth)/n where ‘n’ isthe number of diode connected enhancement-mode transistors betweencurrent mirror 915 and depletion-mode transistor 905.

For example, if the gate of a startup transistor 930 is connected to thethird identical diode connected enhancement-mode low-voltage transistorfrom the bottom, the gate voltage of the startup transistor may be3*(Vpinch−Vth)/n+Vth. Therefore, the startup voltage may be3*(Vpinch−Vth)/n+Vth−Vth=3*(Vpinch−Vth)/n. As a more specific example,in one embodiment where (Vpinch)=40 volts, (Vth)=2 volts where n=6 and(Vstartup)=19 volts.

In other embodiments, startup circuit 155 may generate a referencevoltage signal (Vref). In one embodiment, the circuit that generates(Vref) may be similar to the startup voltage generation circuitdiscussed above. A reference voltage transistor 955 may be connectedbetween two transistors in series of identical diode connectedenhancement-mode low-voltage transistors 910. In one embodiment(Vref)=(Vpinch−Vth)/n.

In further embodiments, a disable pull down transistor 935 may beconnected across the gate to source of startup transistor 930. When thedisable signal is high, startup transistor 930 will be disabled. A pulldown resistor 940 may be connected to the gate of disable transistor 935to prevent false turn on of the disable transistor. In other embodimentsa diode clamp 945 may be connected between the gate and the sourceterminals of startup transistor 930 to ensure that the gate to sourcevoltage capabilities of the startup transistor are not violated duringcircuit operation (i.e., configured as gate overvoltage protectiondevices). In some embodiments, diode clamp 945 may be made with a seriesof diode connected GaN-based enhancement-mode transistors 1050, asillustrated in FIG. 10 .

Now referring to FIG. 11 , UVLO circuit 227 is illustrated in greaterdetail. In some embodiments, UVLO circuit 227 may have a differentialcomparator 1105, a down level shifter 1110 and an inverter 1115. Infurther embodiments, UVLO circuit 227 may use (Vref) and (Iref)generated by startup circuit 155 (see FIG. 9 ) in a differentialcomparator/down level shifter circuit to generate the (LS_UVLO) signalthat feeds into level shift driver circuit 217 (see FIG. 2 ) and lowside transistor driver circuit 120. In some embodiments UVLO circuit 227can also be designed to have asymmetric hysteresis. In furtherembodiments the output of UVLO circuit 227 may be independent ofthreshold voltage. This may be accomplished by choosing a differentialcomparator with a relatively high gain. In one embodiment the gain canbe increased by increasing the value of the current source and the pullup resistors in the differential comparator. In some embodiments thelimit on the current and resistor may be set by (Vref).

In other embodiments voltages (VA) and (VB), 1120 and 1125,respectively, may be proportional to (Vcc) or (Vdd_LS) and (Vref) asdictated by the resistor divider ratio on each input. When (VA)1120>(VB) 1125 the output of the inverting terminal goes to a low state.In one specific embodiment, the low state=(Vth) since the current sourcecreates a source follower configuration. Similarly when (VA) 1120<(VB)1125 the output goes to a high state (Vref). In some embodiments downlevel shifter 1110 may be needed because the low voltage needs to beshifted down by one threshold voltage to ensure that the low input tothe next stage is below (Vth). The down shifted output may be invertedby a simple resistor pull up inverter 1115. The output of inverter 1115is the (LS_UVLO) signal.

Now referring to FIG. 12 , bootstrap capacitor charging circuit 157 isillustrated in greater detail. In one embodiment, bootstrap diode andtransistor circuit 157 may include a parallel connection of a highvoltage diode connected enhancement-mode transistor 1205 and a highvoltage bootstrap transistor 1210. In further embodiments, high voltagediode connected enhancement-mode transistor 1205 and high voltagebootstrap transistor 1210 can be designed to share the same drainfinger. In some embodiments the (BOOTFET_DR) signal may be derived frombootstrap transistor drive circuit 225 (see FIG. 2 ). As discussedabove, high voltage bootstrap transistor 1210 may be turned oncoincident with the turn on of low side transistor 115 (see FIG. 1 ).

Now referring to FIG. 13 , an alternative bootstrap diode and transistorcircuit 1300 may be used in place of bootstrap diode and transistorcircuit 157 discussed above in FIG. 12 . In the embodiment illustratedin FIG. 13 , a depletion-mode device 1305 cascoded by anenhancement-mode low voltage GaN device 1310 may be connected asillustrated in schematic 1300. In another embodiment, a gate ofdepletion-mode device 1305 can be connected to ground to reduce thevoltage stress on cascoded enhancement-mode device 1310, depending uponthe pinch-off voltage of the depletion-mode device.

High Side Device

Now referring to FIG. 14 , high side logic and control circuit 153 isillustrated in greater detail. In one embodiment, high side driver 130receives inputs from first level shift receiver 1410 and high side UVLOcircuit 1415 and sends a (HS_GATE) signal to high side transistor 125(see FIG. 1 ). In yet further embodiments, a pull up trigger circuit1425 is configured to receive the (LSHIFT_1) signal and control pull uptransistor 1435. In some embodiments, second level shift receivercircuit 1420 is configured to control blanking transistor 1440. Both thepull up transistor 1435 and blanking transistor 1440 may be connected inparallel with pull up resistor 1430. Each circuit within high side logicand control circuit 153 is discussed below, and in some cases is shownin more detail in FIGS. 16-20 .

Now referring to FIG. 15 , first level shift receiver 1410 isillustrated in greater detail. In some embodiments, first level shiftreceiver 1410 may convert the (L_SHIFT1) signal to an (LS_HSG) signalthat can be processed by high side transistor driver 130 (see FIG. 14 )to drive high side transistor 125 (see FIG. 1 ). In further embodiments,first level shift receiver 1410 may have three enhancement-modetransistors 1505, 1510, 1515 employed in a multiple level down shifterand a plurality of diode connected transistors 1520 acting as a diodeclamp, as discussed in more detail below.

In one embodiment, first level shift receiver 1410 may down shift the(L_SHIFT1) signal by 3*Vth (e.g., each enhancement-mode transistor 1505,1510, 1515 may have a gate to source voltage close to Vth). In someembodiments the last source follower transistor (e.g., in this casetransistor 1515) may have a three diode connected transistor clamp 1520across its gate to source. In further embodiments this arrangement maybe used because its source voltage can only be as high as (Vdd_HS)(i.e., because its drain is connected to Vdd_HS) while its gate voltagecan be as high as V (L_SHIFT1)−2*Vth. Thus, in some embodiments themaximum gate to source voltage on last source follower transistor 1515may be greater than the maximum rated gate to source voltage of thedevice technology. The output of final source follower transistor 1515is the input to high side transistor drive 130 (see FIG. 1 ), (i.e., theoutput is the LS_HSG signal). In further embodiments fewer or more thanthree source follower transistors may be used. In yet furtherembodiments, fewer or more than three diode connected transistors may beused in clamp 1520.

Now referring to FIG. 16 , second level shift receiver 1420 isillustrated in greater detail. In one embodiment, second level shiftreceiver 1420 may have a down level shift circuit 1605 and an invertercircuit 1610. In some embodiments second level shift receiver 1420 maybe constructed in a similar manner as first level shift receiver 1410(see FIG. 15 ), except the second level shift receiver may have only onedown level shifting circuit (e.g., enhancement-mode transistor 1615) anda follow on inverter circuit 1610. In one embodiment, down level shiftcircuit 1605 may receive the (L_SHIFT2) signal from second level shifttransistor 205 (see FIG. 2 ). In one embodiment, inverter circuit 1610may be driven by the (Vboot) signal, and the gate voltage of the pull uptransistor of the inverter may be used as the (BLANK_FET) signal drivingblanking transistor 1440 (see FIG. 14 ). In some embodiments the voltagemay go from 0 volts in a low state to (Vboot+0.5*(Vboot−Vth)) in a highstate. Similar to first level shift receiver 1410, second level shiftreceiver 1420 may have a diode connected transistor clamp 1620 acrossthe gate to source of source follower transistor 1615. In otherembodiments, clamp 1620 may include fewer or more than three diodeconnected transistors.

Now referring to FIG. 17 , pull up trigger circuit 1425 is illustratedin greater detail. In one embodiment, pull up trigger circuit 1425 mayhave a first inverter 1705, a second inverter 1710, an RC pulsegenerator 1715 and a gate to source clamp 1720. In some embodiments pullup trigger circuit 1425 may receive the (L_SHIFT1) signal as an input,and in response, generate a pulse as soon as the (L_SHIFT1) voltagetransitions to approximately the input threshold of first inverter 1705.The generated pulse may be used as the (PULLUP_FET) signal that drivespull up transistor 1435 (see FIG. 14 ). Second inverter 1710 may bedriven by (Vboot) instead of (Vdd_HS) because pull up transistor 1435gate voltage may need to be larger than the (L_SHIFT1) signal voltage.

Now referring to FIG. 18 , high side UVLO circuit 1415 is illustrated ingreater detail. In one embodiment, high side UVLO circuit 1415 may havedown level shifter 1805, a resistor pull up inverter with asymmetrichysteresis 1810 and a gate to source clamp 1815. In further embodiments,the (HS_UVLO) signal generated by high side UVLO circuit 1415 may aid inpreventing circuit failure by turning off the (HS_GATE) signal generatedby high side drive circuit 130 (see FIG. 14 ) when bootstrap capacitor110 voltage goes below a certain threshold. In some embodiments,bootstrap capacitor 110 voltage (Vboot) (i.e., a floating power supplyvoltage) is measured, and in response, a logic signal is generated andcombined with the output signal (LS_HSG) from first level shift receiver1410 which is then used as the input to the high side gate drive circuit130. More specifically, in this embodiment, for example, the UVLOcircuit is designed to engage when (Vboot) reduces to less than 4*Vthabove switch node (Vsw) 145 voltage. In other embodiments a differentthreshold level may be used.

In further embodiments, high side UVLO circuit 1415 may down shift(Vboot) in down level shifter 1805 and transfer the signal to inverterwith asymmetric hysteresis 1810. The output of inverter with asymmetrichysteresis 1810 may generate the (HS_UVLO) signal which is logicallycombined with the output from the first level shift receiver 1410 toturn off high side transistor 125 (see FIG. 1 ). In some embodiments thehysteresis may be used to reduce the number of self-triggered turn onand turn off events of high side transistor 125 (see FIG. 1 ), that maybe detrimental to the overall performance of half bridge circuit 100.

Now referring to FIG. 19 , high side transistor driver 130 isillustrated in greater detail. High side transistor driver 130 may havea first inverter stage 1905 followed by a high side drive stage 1910.First inverter stage 1905 may invert the down shifted (LS_HSG) signalreceived from level shift 1 receiver 1410 (see FIG. 15 ). Thedownshifted signal may then be sent through high side drive stage 1910.High side drive stage 1910 may generate the (HS_GATE) signal to drivehigh side transistor 125 (see FIG. 1 ). In further embodiments firstinverter stage 1905 may contain a two input NOR gate that may ensurehigh side transistor 125 (see FIG. 1 ) is turned off when the (HS_UVLO)signal is in a high state.

Now referring to FIG. 20 , a reference voltage generation circuit 2000may be used, to generate a high side reference voltage from a supplyrail. Such a circuit maybe placed on the high side GaN device 105 forgenerating internal power supplies which are referenced to the switchnode voltage 145. In some embodiments, circuit 2000 may be similar tostartup circuit 155 in FIG. 9 . One difference in circuit 2000 may bethe addition of a source follower capacitor 2010 connected between firstnode 2011 and second node 2012. In some embodiments, source followercapacitor 2010 may be needed to ensure that a well regulated voltage,which does not fluctuate with dv/dt appearing at the switch node (Vsw)145, develops between the first node 2011 and the second node 2012. Inother embodiments a reference voltage capacitor 2015 may be connectedbetween a source of reference voltage transistor 2055 and second node2012. In some embodiments the drain of the reference voltage transistor2055 may be connected to the (Vboot) node. In some embodiments,reference voltage capacitor 2015 may be needed to ensure that (Vref) iswell regulated and does not respond to high dv/dt conditions at switchnode (Vsw) 145 (see FIG. 1 ). In yet further embodiments, anotherdifference in circuit 2000 may be that second node 2012 may be coupledto a constantly varying voltage, such as switch node (Vsw) 145 (see FIG.1 ), rather than a ground connection through a current sink circuit 915(see FIG. 9 ). In yet further embodiments (Vref) can be used as (Vdd_HS)in the half bridge circuit 100.

Another difference in circuit 2000 may be the addition of a high-voltagediode connected transistor 2025 (i.e., the gate of the transistor iscoupled to the source of the transistor) coupled between depletion-modetransistor 2005 and series of identical diode connected enhancement-modelow-voltage transistors 2020. More specifically, high-voltage diodeconnected transistor 2025 may have source coupled to the source ofdepletion-mode transistor 2005, a drain coupled to first node 2011 and agate coupled to its source. High-voltage diode connected transistor 2025may be used to ensure that source follower capacitor 2010 does notdischarge when the voltage at the top plate of the source followercapacitor rises above (V+). In further embodiments source followercapacitor 2010 may be relatively small and may be integrated on asemiconductor substrate or within an electronic package. Also shown inFIG. 21 is bootstrap capacitor 110 that may be added externally in ahalf bridge circuit.

In some embodiments, shield capacitor 160 (see FIG. 1 ) may be connectedfrom first level shift node 305 (see FIG. 3 ) and second level shiftnode (not shown) to switch node 145 to assist in reducing the falsetriggering discussed above. In some embodiments, the larger the value ofshield capacitor 160, the more immune the circuit will be to falsetriggering effects due to the parasitic capacitance to ground. However,during high side transistor 125 turn off, shield capacitor 160 may bedischarged through pull up resistor 303 (see FIG. 3 ) connected to firstlevel shift node 305. This may significantly slow down high sidetransistor 125 turn off process. In some embodiments this considerationmay be used to set an upper limit on the value of shield capacitor 160.In further embodiments, an overvoltage condition on first level shiftnode 305 (see FIG. 3 ) may be prevented by the use of a clamp circuit161 (see FIG. 1 ) between the first level shift node and switch node145. In some embodiments, clamp circuit 161 maybe composed of a diodeconnected transistor where a drain of the transistor is connected tofirst level shift node 305 (see FIG. 3 ) and a gate and a source areconnected to switch node (Vsw) 145 (see FIG. 1 ). In furtherembodiments, a second shield capacitor and a second clamp circuit may beplaced between the second level shift node and switch node (Vsw) 145(see FIG. 1 ).

Half Bridge Circuit #1 Operation

The following operation sequence for half-bridge circuit 100 is forexample only and other sequences may be used without departing from theinvention. Reference will now be made simultaneously to FIGS. 1, 2 and14 .

In one embodiment, when the (PWM_LS) signal from the controller is high,low side logic, control and level shift circuit 150 sends a high signalto low side transistor driver 120. Low side transistor driver 120 thencommunicates through the (LS_GATE) signal to low side transistor 115 toturn it on. This will set the switch node voltage (Vsw) 145 close to 0volts. When low side transistor 115 turns on, it provides a path forbootstrap capacitor 110 to become charged through bootstrap chargingcircuit 157 which may be connected between (Vcc) and (Vboot). Thecharging path has a parallel combination of a high voltage bootstrapdiode 1205 (see FIG. 12 ) and transistor 1210. The (BOOTFET_DR) signalprovides a drive signal to bootstrap transistor 1210 (see FIG. 12 ) thatprovides a low resistance path for charging bootstrap capacitor 110.

Bootstrap diode 1205 (see FIG. 12 ) may be used to ensure that there isa path for charging bootstrap capacitor 110 during startup when there isno low side transistor 115 gate drive signal (LS_GATE). During this timethe (PWM_HS) signal should be low. If the (PWM_HS) signal isinadvertently turned on (i.e., in a high state) during this time the(STP_HS) signal generated from low side transistor driver 120 willprevent high side transistor 125 from turning on. If the (PWM_LS) signalis turned on while the (PWM_HS) signal is on, the (STP_LS) signalgenerated from level shift driver circuit 217 will prevent low sidetransistor 115 from turning on. Also, in some embodiments the (LS_UVLO)signal may prevent low side transistor 115 and high side transistor 125from turning on when either (Vcc) or (Vdd_LS) goes below a presetthreshold voltage level.

In further embodiments, when the (PWM_LS) signal is low, low side gatesignal (LS_GATE) to low side transistor 115 is also low. During the deadtime between the (PWM_LS) signal low state to the (PWM_HS) high statetransition, an inductive load will force either high side transistor 125or low side transistor 115 to turn on in the synchronous rectifier mode,depending on direction of power flow. If high side transistor 125 turnson during the dead time (e.g., during boost mode operation), switch node(Vsw) 145 voltage may rise close to (V+) 135 (rail voltage).

In some embodiments, a dv/dt condition on switch node 145 (Vsw) may tendto pull first level shift node (LSHIFT_1) 305 (see FIG. 3 ) to a lowstate relative to switch node (Vsw) 145, due to capacitive coupling toground. This may turn on high side gate drive circuit 130 causingunintended triggering of high side transistor 125. In one embodiment,this may result in no dead time which may harm half bridge circuit 100with a shoot through condition. In further embodiments, to prevent thiscondition from occurring, blanking pulse generator 223 may sense theturn off transient of low side transistor 115 and send a pulse to turnon second level shift transistor 205. This may pull the (L_SHIFT2)signal voltage to a low state which then communicates with second levelshift receiver 1420 to generate a blanking pulse signal (B_PULSE) todrive blanking transistor 1440. Blanking transistor 1440 may then act asa pull up to prevent first level shift node (LSHIFT_1) 305 (see FIG. 3 )from going to a low state relative to switch node (Vsw) 145.

In further embodiments, after the dead time, when the (PWM_HS) signalgoes to a high state, level shift driver circuit 217 may send a highsignal to the gate of first level shift transistor 203 (via the L1_DRsignal from level shift driver circuit 217). The high signal will pullfirst level shift node (LSHIFT_1) 305 (see FIG. 3 ) low relative toswitch node (Vsw) 145 which will result in a high signal at the input ofhigh side transistor 125, turning on high side transistor 125. Switchnode voltage (Vsw) 145 will remain close to (V+) 135. In one embodiment,during this time, bootstrap capacitor 110 may discharge through firstlevel shift transistor 203 (which is in an on state during this time).

If high side transistor 125 stays on for a relatively long time (i.e., alarge duty cycle) bootstrap capacitor 110 voltage will go down to a lowenough voltage that it will prevent high side transistor 125 fromturning off when the (PWM_HS) signal goes low. In some embodiments thismay occur because the maximum voltage the (L_SHIFT1) signal can reach is(Vboot) which may be too low to turn off high side transistor 125. Insome embodiments, this situation may be prevented by high side UVLOcircuit 1415 that forcibly turns off high side transistor 125 by sendinga high input to high side gate drive circuit 130 when (Vboot) goes belowa certain level.

In yet further embodiments, when the (PWM_HS) signal goes low, firstlevel shift transistor 203 will also turn off (via the L1_DR signal fromthe level shift driver circuit 217). This will pull first level shiftnode (LSHIFT_1) 305 (see FIG. 3 ) to a high state. However, in someembodiments this process may be relatively slow because the high valuepull up resistor 303 (see FIG. 3 ) (used to reduce power consumption insome embodiments) needs to charge all the capacitances attached to firstlevel shift node (L_SHIFT1) 305 (see FIG. 3 ) including the outputcapacitance (Coss) of first level shift transistor 213 and shieldcapacitor 160. This may increase the turn off delay of high sidetransistor 125. In order to reduce high side transistor 125 turn offdelay, pull up trigger circuit 1425 may be used to sense when firstlevel shift node (L_SHIFT1) 305 (see FIG. 3 ) goes above (Vth). Thiscondition may generate a (PULLUP_FET) signal that is applied to pull uptransistor 1435 which, acting in parallel with pull up resistor 1430,may considerably speed up the pull up of first level shift node(L_SHIFT1) 305 (see FIG. 3 ) voltage, hastening the turn off process.

Half Bridge Circuit #2

Now referring to FIG. 21 , a second embodiment of a half bridge circuit2100 is disclosed. Half bridge circuit 2100 may have the same blockdiagram as circuit 100 illustrated in FIG. 1 , however the level shifttransistors in circuit 2100 may operate with pulsed inputs, rather thana continuous signal, as described in more detail below. In someembodiments, pulsed inputs may result in lower power dissipation,reduced stress on the level shift transistors and reduced switchingtime, as discussed in more detail below.

Continuing to refer to FIG. 21 , one embodiment includes an integratedhalf bridge power conversion circuit 2100 employing a low side GaNdevice 2103, a high side GaN device 2105, a load 2107, a bootstrapcapacitor 2110 and other circuit elements, as discussed in more detailbelow. Some embodiments may also have an external controller (not shownin FIG. 21 ) providing one or more inputs to circuit 2100 to regulatethe operation of the circuit. Circuit 2100 is for illustrative purposesonly and other variants and configurations are within the scope of thisdisclosure.

As further illustrated in FIG. 21 , in one embodiment, integrated halfbridge power conversion circuit 2100 may include a low side circuitdisposed on low side GaN device 2103 that includes a low side transistor2115 having a low side control gate 2117. The low side circuit mayfurther include an integrated low side transistor driver 2120 having anoutput 2123 connected to a low side transistor control gate 2117. Inanother embodiment there may be a high side circuit disposed on highside GaN device 2105 that includes a high side transistor 2125 having ahigh side control gate 2127. The high side circuit may further includean integrated high side transistor driver 2130 having an output 2133connected to high side transistor control gate 2127.

High side transistor 2125 may be used to control the power input intopower conversion circuit 2100 and have a voltage source (V+) 2135(sometimes called a rail voltage) connected to a drain 2137 of the highside transistor. High side transistor 2125 may further have a source2140 that is coupled to a drain 2143 of low side transistor 2115,forming a switch node (Vsw) 2145. Low side transistor 2115 may have asource 2147 connected to ground. In one embodiment, low side transistor2115 and high side transistor 2125 may be enhancement-mode field-effecttransistors. In other embodiments low side transistor 2115 and high sidetransistor 2125 may be any other type of device including, but notlimited to, GaN-based depletion-mode transistors, GaN-baseddepletion-mode transistors connected in series with silicon basedenhancement-mode field-effect transistors having the gate of thedepletion-mode transistor connected to the source of the silicon-basedenhancement-mode transistor, silicon carbide based transistors orsilicon-based transistors.

In some embodiments high side device 2105 and low side device 2103 maybe made from a GaN-based material. In one embodiment the GaN-basedmaterial may include a layer of GaN on a layer of silicon. In furtherembodiments the GaN based material may include, but not limited to, alayer of GaN on a layer of silicon carbide, sapphire or aluminumnitride. In one embodiment the GaN based layer may include, but notlimited to, a composite stack of other III nitrides such as aluminumnitride and indium nitride and III nitride alloys such as AlGaN andInGaN

Low Side Device

Low side device 2103 may have numerous circuits used for the control andoperation of the low side device and high side device 2105. In someembodiments, low side device 2103 may include a low side logic, controland level shift circuit (low side control circuit) 2150 that controlsthe switching of low side transistor 2115 and high side transistor 2125along with other functions, as discussed in more detail below. Low sidedevice 2103 may also include a startup circuit 2155, a bootstrapcapacitor charging circuit 2157 and a shield capacitor 2160, as alsodiscussed in more detail below.

Now referring to FIG. 22 , the circuits within low side control circuit2150 are functionally illustrated. Each circuit within low side controlcircuit 2150 is discussed below, and in some cases is shown in moredetail in FIGS. 23-28 . In one embodiment the primary function of lowside control circuit 2150 may be to receive one or more input signals,such as a PWM signal from a controller, and control the operation of lowside transistor 2115, and high side transistor 2125.

First level shift transistor 2203, may be an “on” pulse level shifttransistor, while second level shift transistor 2215 may be an “off”pulse level shift transistor. In one embodiment, a pulse width modulatedhigh side (PWM_HS) signal from a controller (not shown) may be processedby inverter/buffer 2250 and sent on to an on pulse generator 2260 and anoff pulse generator 2270. On pulse generator 2260 may generate a pulsethat corresponds to a low state to high state transient of the (PWM_HS)signal, thus turning on first level shift transistor 2203 during theduration of the pulse. Off pulse generator 2270 may similarly generate apulse that corresponds to the high state to low state transition of the(PWM_HS) signal, thus turning on second level shift transistor 2205 forthe duration of the off pulse.

First and second level shift transistors 2203, 2205, respectively, mayoperate as pull down transistors in resistor pull up inverter circuits.More specifically, turning on may mean the respective level shift nodevoltages get pulled low relative to switch node (Vsw) 2145 voltage, andturning off may result in the respective level shift nodes assuming the(Vboot) voltage. Since first and second level shift transistors 2203,2215, respectively, are “on” only for the duration of the pulse, thepower dissipation and stress level on these two devices may be less thanhalf bridge circuit 100 illustrated in FIG. 1 .

First and second resistors 2207, 2208, respectively, may be added inseries with the sources of first and second level shift transistors2203, 2215, respectively to limit the gate to source voltage andconsequently the maximum current through the transistors. First andsecond resistors 2207, 2208, respectively, could be smaller than thesource follower resistors in half bridge circuit 100 illustrated in FIG.1 , which may help make the pull down action of first and second levelshift transistors 2203, 2215 faster, reducing the propagation delays tohigh side transistor 2125.

In further embodiments, first and second resistors 2207, 2208,respectively, could be replaced by any form of a current sink. Oneembodiment may connect the source of first and second level shifttransistors 2203, 2205, respectively to a gate to source shorteddepletion-mode device. One embodiment of a depletion-mode transistorformed in a high-voltage GaN technology may be to replace theenhancement-mode gate stack with one of the high-voltage field platemetals superimposed on top of the field dielectric layers. The thicknessof the field dielectric and the work function of the metal may controlthe pinch-off voltage of the stack.

In further embodiments, first and second resistors 2207, 2208,respectively may be replaced by a current sink. In one embodiment areference current (Iref) that is generated by startup circuit 2155 (seeFIG. 21 ) may be used. Both the depletion-mode transistor and currentsink embodiments may result in a significant die area reduction comparedto the resistor option (i.e., because a small depletion transistor wouldsuffice and Iref is already available).

Bootstrap transistor drive circuit 2225 may be similar to bootstraptransistor drive circuit 225 illustrated in FIG. 2 above. Bootstraptransistor drive circuit 2225 may receive input from low side drivecircuit 2220 (see FIG. 22 ) and provide a gate drive signal called(BOOTFET_DR) to the bootstrap transistor in bootstrap capacitor chargingcircuit 2157 (see FIG. 21 ), as discussed in more detail above.

Now referring to FIG. 23 , first level shift transistor 2203 isillustrated along with a pull up resistor 2303 that may be located inhigh side device 2105. In some embodiments, first level shift transistor2203 may operate as a pull down transistor in a resistor pull upinverter similar to first level shift transistor 203 illustrated in FIG.3 . As discussed above, pull up resistor 2303 may be disposed in highside device 2105 (see FIG. 21 ). Second level shift transistor 2215 mayhave a similar configuration. In some embodiments there may be a firstcapacitance between the first output terminal (LS NODE) 2305 and switchnode (Vsw) 2145 (see FIG. 21 ), and a second capacitance between a firstoutput terminal 2305 and ground, where the first capacitance is greaterthan the second capacitance. The first capacitance may be designed suchthat in response to a high dv/dt signal at the switch node (Vsw) 2145(see FIG. 21 ), a large portion of the C*dv/dt current is allowed toconduct through the first capacitance ensuring that the voltage at firstoutput terminal 2305 tracks the voltage at the switch node (Vsw). Ashield capacitor 2160 (see FIG. 21 ) may be configured to act as thefirst capacitor as described above. In further embodiments shieldcapacitor 2160 (see FIG. 21 ) may be used to create capacitance betweenfirst output terminal 2305 and switch node (Vsw) 2145 (see FIG. 21 ) inthe half bridge power conversion circuit 2100. Shield capacitor 2160 mayalso be used to minimize the capacitance between first output terminal2305 and a substrate of the semiconductor device. In further embodimentsshield capacitor 2160 may be constructed on low side GaN device 2103.

Now referring to FIG. 24 , inverter/buffer circuit 2250 is illustratedin greater detail. In one embodiment inverter/buffer circuit 2250 mayhave a first inverter stage 2405 and a first buffer stage 2410. Infurther embodiments, inverter/buffer circuit 2250 may be driven directlyby the (PWM_HS) signal from the controller (not shown). The output offirst inverter stage 2405 may be the input signal (PULSE_ON) to on pulsegenerator 2260 (see FIG. 22 ) while the output of first buffer stage2410 may be an input signal (PULSE_OFF) to off pulse generator 2270.

In some embodiments, an optional (LS_UVLO) signal may be generated bysending a signal generated by UVLO circuit 2227 (see FIG. 22 ) in to aNAND gate disposed in first inverter stage 2405. This circuit may beused to turn off the level shift operation if either (Vcc) or (Vdd_LS)go below a certain reference voltage (or a fraction of the referencevoltage). In further embodiments, inverter/buffer circuit 2250 may alsogenerate a shoot through protection signal (STP_LS1) for low sidetransistor 2115 (see FIG. 21 ) that may be applied to low sidetransistor gate drive circuit 2120. This may turn off low sidetransistor gate drive circuit 2120 (see FIG. 21 ) when the (PWM_HS)signal is high, preventing shoot through.

Now referring to FIG. 25 , on pulse generator 2260 is illustrated ingreater detail. In one embodiment on pulse generator 2260 may have afirst inverter stage 2505, a first buffer stage 2510, an RC pulsegenerator 2515, a second inverter stage 2520 a third inverter stage 2525and a third buffer stage 2530. In further embodiments the (PULSE_ON)signal input from inverter/buffer circuit 2250 (see FIG. 22 ) may befirst inverted and then transformed into an on pulse by RC pulsegenerator 2515 and a square wave generator. The result of this operationis the gate drive signal (LI_DR) that is transmitted to first levelshift transistor 2203 (see FIG. 22 ).

In further embodiments, on pulse generator 2260 may comprise one or morelogic functions, such as for example, a binary or combinatorialfunction. In one embodiment, on pulse generator 2260 may have a multipleinput NOR gate for the (STP_HS) signal. The (STP_HS) signal may have thesame polarity as the (LS_GATE) signal. Therefore, if the (STP_HS) signalis high (corresponding to LS_GATE signal being high) the on pulse maynot be generated because first inverter circuit 2505 in FIG. 25 will bepulled low which will deactivate pulse generator 2515.

In further embodiments, RC pulse generator 2515 may include a clampdiode (not shown). The clamp diode may be added to ensure that RC pulsegenerator 2515 works for very small duty cycles for the (PWM_LS) signal.In some embodiments, on pulse generator 2260 may be configured toreceive input pulses in a range of 2 nanoseconds to 20 microseconds andto transmit pulses of substantially constant duration within the range.In one embodiment the clamp diode may turn on and short out a resistorin RC pulse generator 2515 (providing a very small capacitor dischargetime) if the voltage across the clamp diode becomes larger than (Vth).This may significantly improve the maximum duty cycle of operation (withrespect to the PWM_HS signal) of pulse generator circuit 2260.

Now referring to FIG. 26 , off pulse generator 2270 is illustrated ingreater detail. In one embodiment off pulse generator 2270 may have anRC pulse generator 2603, a first inverter stage 2605, a second inverterstage 2610 and a first buffer stage 2615. In further embodiments, offpulse generator 2270 may receive an input signal (PULSE_OFF) frominverter/buffer circuit 2250 (see FIG. 22 ) that may be subsequentlycommunicated to RC pulse generator 2603.

In further embodiments the pulse from RC pulse generator 2603 is sentthrough first inverter stage 2605, second inverter stage 2610 and bufferstage 2615. The pulse may then be sent as the (L2_DR) signal to secondlevel shift transistor 2215 (see FIG. 22 ). A clamp diode may also beincluded in off pulse generator 2270. In some embodiments, the operatingprinciple may be similar to the operating principle discussed above withregard to on pulse generator 2260 (see FIG. 25 ). Such operatingprinciples may ensure that off pulse generator 2270 operates for verylow on times of high side transistor 2125 (see FIG. 21 ) (i.e. thecircuit will operate for relatively small duty cycles). In someembodiments, off pulse generator 2270 may be configured to receive inputpulses in a range of 2 nanoseconds to 20 microseconds and to transmitpulses of substantially constant duration within the range. In furtherembodiments an off level shift pulse can be shortened by an on inputpulse to enable an off time of less than 50 nanoseconds on high sidetransistor 2125.

In some embodiments, RC pulse generator 2603 may include a capacitorconnected with a resistor divider network. The output from the resistormay be a signal (INV) that is sent to an inverter 2275 (see FIG. 22 )that generates a shoot through protection signal (STP_LS2) transmittedto low side driver circuit 2220. In further embodiments, off pulsegenerator 2270 may comprise one or more logic functions, such as forexample, a binary or combinatorial function. In one embodiment the(STP_LS2) signal is sent to a NAND logic circuit within low side drivercircuit 2220, similar to the (STP_LS1) signal. In some embodiments,these signals may be used to ensure that during the duration of the offpulse signal (PULSE_OFF), low side transistor 2115 (see FIG. 21 ) doesnot turn on (i.e., because high side transistor 2125 turns off duringthe off pulse). In some embodiments this methodology may be useful tocompensate for a turn off propagation delay (i.e., the PULSE_OFF signalmay enable shoot through protection), ensuring that low side transistor2115 will only turn on after high side transistor 2125 gate completelyturns off.

In further embodiments, a blanking pulse can be level shifted to highside device 2105 using second level shift transistor 2215. To accomplishthis, a blanking pulse may be sent into a NOR input into first inverterstage 2605. The blanking pulse may be used to inhibit false triggeringdue to high dv/dt conditions at switch node Vsw 2145 (see FIG. 20 ). Insome embodiments no blanking pulse may be used to filter dv/dt inducedor other unwanted level shift output pulses.

Now referring to FIG. 27 , blanking pulse generator 2223 is illustratedin greater detail. In one embodiment, blanking pulse generator 2223 maybe a more simple design than used in half bridge circuit 100 illustratedin FIG. 1 because the square wave pulse generator is already part of offpulse generator 2270. In one embodiment the (LS_GATE) signal is fed asthe input to blanking pulse generator 2223 from low side gate drivecircuit 2220 (see FIG. 22 ). This signal may be inverted and then sentthrough an RC pulse generator to generate a positive going pulse. Insome embodiments, an inverted signal may be used because the pulse needsto correspond to the falling edge of the (LS_GATE) signal. The output ofthis may be used as the blanking pulse input (B_PULSE) to off pulsegenerator 2270.

Now referring to FIG. 28 , low side transistor drive circuit 2220 isillustrated in greater detail. In one embodiment low side transistordrive circuit 2220 may have a first inverter stage 2805, a first bufferstage 2810, a second inverter stage 2815, a second buffer stage 2820 anda third buffer stage 2825. In some embodiments two inverter/bufferstages may be used because the input to the gate of low side transistor2115 is synchronous with the (PWM_LS) signal. Thus, in some embodimentsa (PWM_LS) high state may correspond to a (LS_GATE) high state and viceversa.

In further embodiments, low side transistor drive circuit 2220 may alsoinclude an asymmetric hysteresis using a resistor divider with atransistor pull down similar to the scheme described in 120 (see FIG. 8). In one embodiment low side transistor drive circuit 2220 includesmultiple input NAND gates for the (STP_LS1) and (STP_LS2) (shoot throughprevention on low side transistor 2115) signals. The (STP_LS1) and(STP_LS2) signals may ensure that low side transistor drive circuit 2220(see FIG. 22 ) does not communicate with low side transistor 2115 (seeFIG. 21 ) when high side transistor 2125 is on. This technique may beused to avoid the possibility of shoot-through. Other embodiments mayinclude NAND gates (similar to the ones employed above in FIG. 28 ) forthe (LS_UVLO) signal. One embodiment may include a turn off delayresistor in series with the gate of the final pull down transistor. Thismay be used to ensure the bootstrap transistor is turned off before lowside transistor 2115 turns off.

In further embodiments, low side device 2103 (see FIG. 21 ) may alsoinclude a startup circuit 2155, bootstrap capacitor charging circuit2157, a shield capacitor 2160, and a UVLO circuit 2227 that may besimilar to startup circuit 155, bootstrap capacitor charging circuit157, shield capacitor 160 and UVLO circuit 227, respectively, asdiscussed above.

High Side Device

Now referring to FIG. 29 , high side logic and control circuit 2153 andhow it interacts with high side transistor driver 2130 is illustrated ingreater detail. In some embodiments, high side logic and control circuit2153 may operate in similar ways as high side logic and control circuit153, discussed above in FIG. 15 . In further embodiments, high sidelogic and control circuit 2153 may operate in different ways, asdiscussed in more detail below.

In one embodiment, level shift 1 receiver circuit 2910 receives an(L_SHIFT1) signal from first level shift transistor 2203 (see FIG. 22 )that receives an on pulse at the low state to high state transition ofthe (PWM_HS) signal, as discussed above. In response, level shift 1receiver circuit 2910 drives a gate of pull up transistor 2960 (e.g., insome embodiments a low-voltage enhancement-mode GaN transistor). Infurther embodiments, pull up transistor 2960 may then pull up a statestoring capacitor 2955 voltage to a value close to (Vdd_HS) with respectto switch node (Vsw) 2145 voltage. The voltage on a state storingcapacitor 2955 may then be transferred to high side transistor driver2130 and on to the gate of high side transistor gate 2127 (see FIG. 21 )to turn on high side transistor 2125. In some embodiments state storingcapacitor 2955 may be a latching storage logic circuit configured tochange state in response to a first pulsed input signal and to changestate in response to a second pulsed input signal. In furtherembodiments, state storing capacitor 2955 may be replaced by any type ofa latching circuit such as, but not limited to an RS flip-flop.

In further embodiments, during this time, level shift 2 receiver circuit2920 may maintain pull down transistor 2965 (e.g., in some embodiments alow-voltage enhancement-mode GaN transistor) in an off state. This maycut off any discharge path for state storing capacitor 2955. Thus, insome embodiments, state storing capacitor 2955 may have a relativelysmall charging time constant and a relatively large discharge timeconstant.

Similarly, level shift 2 receiver 2920 may receive an (L_SHIFT2) signalfrom second level shift transistor 2215 (see FIG. 22 ) that receives anoff pulse at the high state to low state transition of the (PWM_HS)signal, as discussed above. In response, level shift 2 receiver circuit2920 drives a gate of pull down transistor 2965 (e.g., in someembodiments a low-voltage enhancement-mode GaN transistor). In furtherembodiments, pull down transistor 2965 may then pull down (i.e.,discharge) state storing capacitor 2955 voltage to a value close toswitch node (Vsw) 2145, that may consequently turn off high sidetransistor 2125 through high side transistor driver 2130.

Continuing to refer to FIG. 29 , first and second shield capacitors2970, 2975, respectively, may be connected from (L_SHIFT1) and(L_SHIFT2) nodes to help prevent false triggering during high dv/dtconditions at switch node (Vsw) 2145 (see FIG. 21 ). In furtherembodiments there may also be a clamp diode between the (L_SHIFT1) and(L_SHIFT2) nodes and the switch node (Vsw) 2145 (see FIG. 21 ). This mayensure that the potential difference between switch node (Vsw) 2145 (seeFIG. 21 ) and the (L_SHIFT1) and (L_SHIFT2) nodes never goes above(Vth). This may be used to create a relatively fast turn on and turn offfor high side transistor 2125 (see FIG. 21 ).

Now referring to FIG. 30 , level shift 1 receiver 2910 is illustrated ingreater detail. In one embodiment level shift 1 receiver 2910 mayinclude a down level shifter 3005, a first inverter 3010, a secondinverter 3015, a first buffer 3020, a third inverter 3025, a secondbuffer 3030 and a third buffer 3135. In some embodiments, level shift 1receiver 2910 down shifts (i.e., modulates) the (L_SHIFT1) signal by avoltage of 3*Vth (e.g., using three enhancement-mode transistors whereeach may have a gate to source voltage close to Vth). In otherembodiments a fewer or more downshift transistors may be used.

In further embodiments, the last source follower transistor may have athree diode connected transistor clamp across its gate to its source. Insome embodiments this configuration may be used because its sourcevoltage can only be as high as (Vdd_HS) (i.e., because its drain isconnected to Vdd_HS) while its gate voltage can be as high as V(L_SHIFT1)−2*Vth. Thus, in some embodiments the maximum gate to sourcevoltage on the final source follower transistor can be greater than themaximum rated gate to source voltage in the technology.

In further embodiments, first inverter 3010 may also have a NOR Gate forthe high side under voltage lock out using the (UV_LS1) signal generatedby high side UVLO circuit 2915. In one embodiment, an output of levelshift 1 receiver 2910 (see FIG. 29 ) may be a (PU_FET) signal that iscommunicated to a gate of pull up transistor 2960 (see FIG. 29 ). Thissignal may have a voltage that goes from 0 volts in a low state to(Vdd_HS)+(Vdd_HS−Vth) in a high state. This voltage may remain on forthe duration of the on pulse.

Now referring to FIG. 31 , level shift 2 receiver 2920 is illustrated ingreater detail. In one embodiment level shift 2 receiver 2920 may besimilar to level shift 1 receiver 2910 discussed above. In furtherembodiments level shift 2 receiver 2920 may include a blanking pulsegenerator 3105, a down level shifter 3110, a first inverter 3115, asecond inverter 3120, a first buffer 3125, an third inverter 3130, asecond buffer 3135 and a third buffer 3140. In one embodiment, blankingpulse generator 3105 may be used in addition to a 3*Vth down levelshifter 3110 and multiple inverter/buffer stages.

In other embodiments different configurations may be used. In someembodiments, this particular configuration may be useful when levelshift 2 receiver 2920 doubles as a high side transistor 2125 (see FIG.21 ) turn off as well as a blanking transistor 2940 (see FIG. 29 ) drivefor better dv/dt immunity. In some embodiments, blanking pulse generator3105 may be identical to level shift 2 receiver 1520 illustrated in FIG.17 . In one embodiment level shift 2 receiver 2920 (see FIG. 29 ) mayreceive (L_SHIFT2) and (UV_LS2) signals and in response, transmit a(PD_FET) signal to pull down transistor 2965. In further embodiments,first inverter 3115 may have a two input NAND gate for the (UV_LS2)signal from high side UVLO circuit 2915 (see FIG. 29 ).

Now referring to FIG. 32 , high side UVLO circuit 2915 is illustrated ingreater detail. In one embodiment high side UVLO circuit 2915 mayinclude a down level shifter 3205 and a resistor pull up inverter stage3210. In some embodiments, high side UVLO circuit 2915 may be configuredto prevent circuit failure by turning off the (HS_GATE) signal to highside transistor 2125 (see FIG. 21 ) when bootstrap capacitor 2110voltage goes below a certain threshold. In one example embodiment highside UVLO circuit 2915 is designed to engage when (Vboot) reduces to avalue less than 4*Vth below switch node (Vsw) 2145 voltage. In anotherembodiment the output of down level shifter 3205 may be a (UV_LS2)signal transmitted to second level shift receiver 2920 and the output ofresistor pull up inverter stage 3210 may be an (UV_LS1) signal that istransmitted to first level shift receiver 2910.

As discussed below, in some embodiments high side UVLO circuit 2915 maybe different from high side UVLO circuit 1415 for half bridge circuit100 discussed above in FIGS. 14 and 18 , respectively. In oneembodiment, the (Vboot) signal may be down shifted by 3*Vth andtransferred to resistor pull up inverter stage 3210. In furtherembodiments, since level shift 2 receiver circuit 2920 (see FIG. 29 )controls the turn off process based on high side transistor 2125 (seeFIG. 21 ), directly applying a 3*Vth down shifted output to the NANDgate at the input of level shift 2 receiver circuit 2920 will engage theunder voltage lock out.

However, in some embodiments, because the bootstrap voltage may be toolow this may also keep pull up transistor 2960 (see FIG. 29 ) on. Insome embodiments, this may result in a conflict. While level shift 2receiver circuit 2920 (see FIG. 29 ) tries to keep high side transistor2125 (see FIG. 21 ) off, level shift 1 receiver circuit 2910 may try toturn the high side transistor on. In order to avoid this situation, someembodiments may invert the output of the 3*Vth down shifted signal fromhigh side UVLO circuit 2915 (see FIG. 29 ) and send it to a NOR input onlevel shift 1 receiver circuit 2910. This may ensure that level shift 1receiver circuit 2910 does not interfere with the UVLO induced turn offprocess.

Now referring to FIG. 33 , high side transistor driver 2130 isillustrated in greater detail. In one embodiment high side transistordriver 2130 may include a first inverter 3305, a first buffer 3310, asecond inverter 3315, a second buffer 3320 and a third buffer 3325. Insome embodiments high side transistor driver 2130 may be a more basicdesign than high side transistor driver 130 employed in half bridgecircuit 100 illustrated in FIG. 1 . In one embodiment, high sidetransistor driver 2130 receives an (S CAP) signal from state storagecapacitor 2955 (see FIG. 29 ) and delivers a corresponding drive(HS_GATE) signal to high side transistor 2125 (see FIG. 21 ). Morespecifically, when the (S CAP) signal is in a high state, the (HS_GATE)signal is in a high state and vice versa.

Half Bridge Circuit #2 Operation

The following operation sequence for half-bridge circuit 2100 (see FIG.21 ) is for example only and other sequences may be used withoutdeparting from the invention. Reference will now be made simultaneouslyto FIGS. 21, 22 and 29 .

In one embodiment, when the (PWM_LS) signal is in a high state, low sidelogic, control and level shift circuit 2150 may send a high signal tolow side transistor driver 2120 which then communicates that signal tolow side transistor 2115 to turn it on. This may set switch node (Vsw)2145 voltage close to 0 volts. In further embodiments, when low sidetransistor 2115 turns on it may provide a path for bootstrap capacitor2110 to charge. The charging path may have a parallel combination of ahigh-voltage bootstrap diode and transistor.

In some embodiments, bootstrap transistor drive circuit 2225 may providea drive signal (BOOTFET_DR) to the bootstrap transistor that provides alow resistance path for charging bootstrap capacitor 2110. In oneembodiment, the bootstrap diode may ensure that there is a path forcharging bootstrap capacitor 2110 during startup when there is no lowside gate drive signal (LS_GATE). During this time the (PWM_HS) signalshould be in a low state. If the (PWM_HS) signal is inadvertently turnedon during this time, the (STP_HS) signal generated from low side drivercircuit 2220 may prevent high side transistor 2125 from turning on. Ifthe (PWM_LS) signal is turned on while the (PWM_HS) signal is on, thenthe (STP_LS1) and (STP_LS2) signals generated from inverter/buffer 2250and inverter 2275, respectively will prevent low side transistor 2115from turning on. In addition, in some embodiments the (LS_UVLO) signalmay prevent low side gate 2117 and high side gate 2127 from turning onwhen either (Vcc) or (Vdd_LS) go below a predetermined voltage level.

Conversely, in some embodiments when the (PWM_LS) signal is in a lowstate, the (LS_GATE) signal to low side transistor 2115 may also be in alow state. In some embodiments, during the dead time between the(PWM_LS) low signal and the (PWM_HS) high signal transition, theinductive load may force either high side transistor 2125 or low sidetransistor 2115 to turn on in the synchronous rectifier mode, dependingon the direction of power flow. If high side transistor 2125 turns onduring the dead time (e.g., in a boost mode), switch node (Vsw) 2145voltage may rise close to (V+) 2135 (i.e., the rail voltage). This dv/dtcondition on switch node (Vsw) 2145 may tend to pull the (L_SHIFT1) nodeto a low state relative to the switch node (i.e., because of capacitivecoupling to ground) which may turn on high side transistor driver 2130causing unintended conduction of high side transistor 2125. Thiscondition may negate the dead time, causing shoot through.

In some embodiments this condition may be prevented by using blankingpulse generator 2223 to sense the turn off transient of low sidetransistor 2115 and send a pulse to turn on second level shifttransistor 2205. This may pull the (L_SHIFT2) signal to a low statewhich may then communicate with level shift 2 receiver circuit 2920 togenerate a blanking pulse to drive blanking transistor 2940. In oneembodiment, blanking transistor 2940 may act as a pull up to prevent the(L_SHIFT1) signal from going to a low state relative to switch node(Vsw) 2145.

In further embodiments, after the dead time when the (PWM_HS) signaltransitions from a low state to a high state, an on pulse may begenerated by on pulse generator 2260. This may pull the (L_SHIFT1) nodevoltage low for a brief period of time. In further embodiments thissignal may be inverted by level shift 1 receiver circuit 2910 and abrief high signal will be sent to pull up transistor 2960 that willcharge state storage capacitor 2955 to a high state. This may result ina corresponding high signal at the input of high side transistor driver2130 which will turn on high side transistor 2125. Switch node (Vsw)2145 voltage may remain close to (V+) 2135 (i.e., the rail voltage).State storing capacitor 2955 voltage may remain at a high state duringthis time because there is no discharge path.

In yet further embodiments, during the on pulse, bootstrap capacitor2110 may discharge through first level shift transistor 2203. However,since the time period is relatively short, bootstrap capacitor 2110 maynot discharge as much as it would if first level shift transistor 2203was on during the entire duration of the (PWM_HS) signal (as was thecase in half bridge circuit 100 in FIG. 1 ). More specifically, in someembodiments this may result in the switching frequency at which the UVLOengages to be a relatively lower value than in half bridge circuit 100in FIG. 1 .

In some embodiments, when the (PWM_HS) signal transitions from a highstate to a low state, an off pulse may be generated by off pulsegenerator 2270. This may pull the (L_SHIFT2) node voltage low for abrief period of time. This signal may be inverted by level shift 2receiver circuit 2920 and a brief high state signal may be sent to pulldown transistor 2965 that will discharge state storing capacitor 2955 toa low state. This will result in a low signal at the input of high sidetransistor driver 2130 that will turn off high side transistor 2125. Infurther embodiments, state storing capacitor 2955 voltage may remain ata low state during this time because it has no discharge path.

In one embodiment, since the turn off process in circuit 2100 does notinvolve charging level shift node capacitors through a high value pullup resistor, the turn off times may be relatively shorter than in halfbridge circuit 100 in FIG. 1 . In further embodiments, high sidetransistor 2125 turn on and turn off processes may be controlled by theturn on of substantially similar level shift transistors 2203, 2205,therefore the turn on and turn off propagation delays may besubstantially similar. This may result in embodiments that have no needfor a pull up trigger circuit and/or a pull up transistor as were bothused in half bridge circuit 100 in FIG. 1 .

ESD Circuits

Now referring to FIG. 34 , in some embodiments, one or more pins (i.e.,connections from a semiconductor device within an electronic package toan external terminal on the electronic package) may employ anelectro-static discharge (ESD) clamp circuit to protect the circuit. Thefollowing embodiments illustrate ESD clamp circuits that may be used onone or more pins in one or more embodiments disclosed herein, as well asother embodiments that may require ESD protection. In furtherembodiments, the ESD clamp circuits disclosed herein may be employed onGaN-based devices.

One embodiment of an electro-static discharge (ESD) clamp circuit 3400is illustrated. ESD clamp circuit 3400 may have a configurationemploying one or more source follower stages 3405 made fromenhancement-mode transistors. Each source follower stage 3405 may have agate 3406 connected to a source 3407 of an adjacent source followerstage. In the embodiment illustrated in FIG. 34 , four source followerstages 3405 are employed, however in other embodiments fewer or more maybe used. Resistors 3410 are coupled to sources 3407 of source followerstages 3405.

An ESD transistor 3415 is coupled to one or more source follower stages3405 and may be configured to conduct a current greater than 500 mA whenexposed to an overvoltage pulse, as discussed below. Resistors 3410 aredisposed between source 3420 of ESD transistor 3415 and each source 3407of source follower stages 3405. Drains 3408 of source follower stages3405 are connected to drain 3425 of ESD transistor 3415. Source 3407 ofthe last source follower stage is coupled to gate 3430 of ESD transistor3415.

In one embodiment, a turn on voltage of ESD clamp circuit 3400 can beset by the total number of source follower stages 3405. However, sincethe last source follower stage is a transistor with a certain drain 3408to source 3407 voltage and gate 3406 to source voltage the currentthrough the final resistor 3410 may be relatively large and may resultin a larger gate 3430 to source 3420 voltage across ESD transistor 3415.This condition may result in a relatively large ESD current capabilityand in some embodiments an improved leakage performance compared toother ESD circuit configurations.

In further embodiments, ESD clamp circuit 3400 may have a plurality ofdegrees of freedom with regard to transistor sizes and resistor values.In some embodiments ESD clamp circuit 3400 may be able to be madesmaller than other ESD circuit configurations. In other embodiments, theperformance of ESD clamp circuit 3400 may be improved by incrementallyincreasing the size of source follower stages 3405 as they get closer toESD transistor 3415. In further embodiments, resistors 3410 can bereplaced by depletion-mode transistors, reference current sinks orreference current sources, for example.

Now referring to FIG. 35 an embodiment similar to ESD clamp circuit 3400in FIG. 34 is illustrated, however ESD clamp circuit 3500 may haveresistors in a different configuration, as discussed in more detailbelow. ESD clamp circuit 3500 may have a configuration employing one ormore source follower stages 3505 made from one or more enhancement-modetransistors. Each source follower stage 3505 may have a gate 3506connected to a source 3507 of an adjacent source follower stage. In theembodiment illustrated in FIG. 35 , four source follower stages 3505 areemployed, however in other embodiments fewer or more may be used.Resistors 3510 are coupled between sources 3507 of adjacent sourcefollower stages 3505. An ESD transistor 3515 is coupled to sourcefollower stages 3505 with resistor 3510 disposed between source 3520 ofESD transistor 3515 and source 3507 of a source follower stage 3505.Drains 3508 of source follower stages 3505 may be coupled together andto drain 3525 of ESD transistor 3515.

Electronic Packaging

Now referring to FIGS. 36 and 37 , in some embodiments, one or moresemiconductor devices may be disposed in one or more electronicpackages. Myriad packaging configurations and types of electronicpackages are available and are within the scope of this disclosure. FIG.36 illustrates one example of what is known as a quad-flat no-leadelectronic package with two semiconductor devices within it.

Electronic package 3600 may have a package base 3610 that has one ormore die pads 3615 surrounded by one or more terminals 3620. In someembodiments package base 3610 may comprise a leadframe while in otherembodiments it may comprise an organic printed circuit board, a ceramiccircuit or another material.

In the embodiment depicted in FIG. 36 , a first device 3620 is mountedto a first die pad 3615 and a second device 3625 is mounted to a seconddie pad 3627. In another embodiment one or more of first and seconddevices 3620, 3625, respectively may be mounted on an insulator (notshown) that is mounted to package base 3610. In one embodiment theinsulator may be a ceramic or other non-electrically conductivematerial. First and second devices 3620, 3625, respectively areelectrically coupled to terminals 3640 with wire bonds 3630 or any othertype of electrical interconnect such as, for example, flip-chip bumps orcolumns that may be used in a flip-chip application. Wirebonds 3630 mayextend between device bond pads 3635 to terminals 3640, and in somecases to die pads 3615, 3627 and in other cases to device bond pads 3635on an adjacent device.

Now referring to FIG. 37 , an isometric view of electronic package 3600is shown. Terminals 3640 and die attach pads 3615 and 3627 may bedisposed on an external surface and configured to attach to a printedcircuit board or other device. In further embodiments, terminals 3640and die attach pads 3615 and 3627 may only be accessible within theinside of electronic package 3600 and other connections may be disposedon the outside of the electronic package. More specifically, someembodiments may have internal electrical routing and there may not be aone to one correlation between internal and external connections.

In further embodiments first and second devices 3620, 3625, respectively(see FIG. 36 ) and a top surface of package base 3610 may beencapsulated by a non-electrically conductive material, such as forexample, a molding compound. Myriad other electronic packages may beused such as, but not limited to, SOIC's, DIPS, MCM's and others.Further, in some embodiments each device may be in a separate electronicpackage while other embodiments may have two or more electronic deviceswithin a single package. Other embodiments may have one or more passivedevices within one or more electronic packages.

In the foregoing specification, embodiments of the invention have beendescribed with reference to numerous specific details that may vary fromimplementation to implementation. The specification and drawings are,accordingly, to be regarded in an illustrative rather than a restrictivesense. The sole and exclusive indicator of the scope of the invention,and what is intended by the applicants to be the scope of the invention,is the literal and equivalent scope of the set of claims that issue fromthis application, in the specific form in which such claims issue,including any subsequent correction.

1. (canceled)
 2. A circuit comprising: a first transistor having a firstgate terminal, a first source terminal and a first drain terminal; and asecond transistor having a second gate terminal, a second sourceterminal and a second drain terminal; wherein the first drain terminalis coupled to the second source terminal.
 3. The circuit of claim 2,wherein the first transistor is a gallium nitride (GaN)-basedtransistor.
 4. The circuit of claim 3, wherein the first transistor isan enhancement-mode GaN-based transistor.
 5. The circuit of claim 4,wherein the second transistor is a GaN-based transistor.
 6. The circuitof claim 5, wherein the second transistor is an enhancement-modeGaN-based transistor.
 7. The circuit of claim 2, wherein the firsttransistor is a silicon-based transistor.
 8. The circuit of claim 7,wherein the second transistor is a silicon-based transistor.
 9. Thecircuit of claim 5, wherein the first transistor is disposed on a firstGaN-based substrate and the second transistor is disposed on a secondGaN-based substrate.
 10. The circuit of claim 8, wherein the firsttransistor is disposed on a first silicon-based substrate and the secondtransistor is disposed on a second silicon-based substrate.
 2. Thecircuit of claim 2, further comprising a first gate driver circuitcoupled to the first gate terminal, wherein the first gate drivercircuit is arranged to receive a first input signal that is referencedto ground and control a conductivity state of the first transistor. 12.The circuit of claim 11, further comprising a second gate driver circuitcoupled to the second gate terminal, wherein the second gate drivercircuit is arranged to receive a second input signal that is referencedto a floating voltage and control conductivity state of the secondtransistor.
 13. An electronic component comprising: a package base; oneor more semiconductor dies secured to the package base and comprising: afirst transistor having a first gate terminal, a first source terminaland a first drain terminal; and a second transistor having a second gateterminal, a second source terminal and a second drain terminal; whereinthe first drain terminal is coupled to the second source terminal. 14.The electronic component of claim 13, wherein the first transistor is agallium nitride (GaN)-based transistor.
 15. The electronic component ofclaim 14, wherein the first transistor is an enhancement-mode GaN-basedtransistor.
 16. The electronic component of claim 15, wherein the secondtransistor is a GaN-based transistor.
 17. The electronic component ofclaim 16, wherein the second transistor is an enhancement-mode GaN-basedtransistor.
 18. The electronic component of claim 13, wherein the firsttransistor is a silicon-based transistor.
 19. The electronic componentof claim 18, wherein the second transistor is a silicon-basedtransistor.
 20. The electronic component of claim 16, wherein the one ormore semiconductor dies comprise a first and second GaN-based dies, andwherein the first transistor is disposed on the first GaN-based die andthe second transistor is disposed on the second GaN-based die.
 21. Theelectronic component of claim 19, wherein the one or more semiconductordies comprise a first and second silicon-based dies, and wherein thefirst transistor is disposed on the first silicon-based die and thesecond transistor is disposed on the second silicon-based die.